Frequency stabilized microwave source using an IQ mixer to detect amplitude modulation of the reflected signal

ABSTRACT

An IQ mixer is used in a Pound-stabilized microwave source to detect amplitude modulation of the signal reflected from the reference resonator. By properly configuring the IQ mixer so that the LO and RF inputs are maintained in quadrature at the Q mixer, hence in-phase at the I mixer, lower levels of amplitude modulation may be detected at lower modulation frequencies compatible with optimal choices of resonator coupling and maximal phase to amplitude conversion.

BACKGROUND Field

This disclosure relates to frequency stabilized microwave sources thatuse “Pound stabilization” or “Pound-servo” techniques, and moreparticularly to the use of an IQ mixer to detect the amplitudemodulation of the reflected signal.

Description of the Related Art

Stable microwave frequency sources are used in applications such asradar and communications systems. The most stable microwave sources, forexample, hydrogen masers or cryogenic sapphire oscillators, findapplication as the time base for accurate clocks. Such sources andclocks are bulky and unsuitable for portable devices which may requiresynchronization even when separated by large distances. Therefore, manysuch devices will synchronize by utilizing the time information providedby GPS signals. However, GPS reception may be unavailable for someperiods and there is a need for small, low power, oscillators withsufficient stability to maintaining synchronization during such periods.

A simple oscillator may comprise a loop that includes at least aresonator, an amplifier, and a phase shifter. Together these may satisfythe oscillation conditions of gain and phase shift around the loop beinggreater than 1 and a multiple of 2π, respectively. A resonator with ahighly stable center frequency does not immediately lead to a highlystable oscillation frequency if the phase shift through the amplifierand phase shifter are not also highly stable.

R. V. Pound, “Frequency Stabilization of Microwave Oscillators,” inProceedings of the IRE, vol. 35, no. 12, pp. 1405-1415, December 1947described a technique which utilized the reflection of the microwaveoscillator signal from a reference resonator to improve oscillatorstability. In U.S. Pat. No. 2,681,998 Pound combined this technique withmodulation of the oscillator signal to avoid limitations induced byflicker noise in microwave detectors. It has become common to refer tomicrowave oscillator stabilization techniques involving reflection of amodulated signal from a resonator as “Pound-stabilization” or“Pound-servo” techniques. More recently a single resonator is oftenutilized as both a component of a microwave oscillator loop as well asthe stabilizing element used to improve the stability of the oscillatorloop. See N. Luiten, A. G. Mann, N. J. McDonald and D. G. Blair, “Latestresults of the U.W.A. cryogenic sapphire oscillator,” Proceedings of the1995 IEEE International Frequency Control Symrposiuni (49th AnnualSymposium), 1995, pp. 433-437 and N. Luiten, A. G. Mann, M. E. Costa andD. G. Blair, “Cryogenic sapphire resonator-oscillator with exceptionalstability: an update,” Proceedings of IEEE 48th Annual Symposium onFrequency Control, 1994, pp. 441-446.

Referring now to FIG. 1 , a frequency stabilized microwave source 100uses the Pound-stabilization technique to stabilize a microwaveoscillator i.e., a voltage controlled oscillator (VCO) 102 against areference resonator 104 to generate a frequency-stabilized output signal106 at output port 108. The VCO generates an output signal 110, aportion of which is taken as frequency-stabilized output signal 106. Thesignal frequency can be tuned by a Pound-servo voltage 112 applied tothe VCO's tuning port 114. Output signal 110 is passed through a phaseshifter 116 and a circulator 118 before impinging on reference resonator104. The frequency of the VCO 102 must be tunable to match the centerfrequency of the resonator, f_(Res). Typically, the resonator may be acavity resonator which may include a dielectric element, although thetechnique may be applied using any resonator with a suitable centerfrequency and stability.

A portion of the output signal 110 impinging on the resonator will bereflected in accordance with well-known properties of resonators asreflected signal 120. This reflected energy is directed by circulator118 onto an amplitude detector 122 which is most commonly a simple diodedetector. A modulation source 124 generates a modulation signal 125 at amodulation frequency, f_(m) to apply phase modulation to output signal110 impinging on the resonator by way of the phase shifter 116. As willbe described in more detail below, the phase modulation may lead toamplitude modulation in the reflected signal 120 seen by the amplitudedetector 122. Any such amplitude modulation will cause a fluctuatingsignal on the detector output, said fluctuation having the samemodulation frequency f_(m).

A synchronous detector 126 such as a synchronous rectifier mixes theoutput of the amplitude detector 122 with the modulation signal 125 toconvert the fluctuation in amplitude detector output at modulationfrequency f_(m) into a DC frequency error voltage 128 representing themagnitude of the fluctuation. This DC frequency error voltage is thenamplified and preferably integrated via integrator 130 to providePound-servo gain to produce Pound-servo voltage 112 that is fed to thetuning port 114 of VCO 102. The synchronous rectifier may also be knownin the art by terms such as a synchronous detector, lock-in detector,lock-in amplifier or lock-in mixer.

To understand the operation of the frequency stabilized microwavesource, consider an incident spectrum 200 of the signal impinging on thereference resonator as shown in FIG. 2A and a variation of a complexreflection coefficient 202 of the reference resonator with frequency asseparate magnitude and phase plots 204 and 206, respectively as shown inFIG. 2B. For the purposes of explanation, the VCO frequency, f_(c), ofcarrier 208 is intentionally misaligned with the center frequency of theresonator, f_(c)≠f_(Res). The first pair of phase modulation sidebands210 and 212 are located symmetrically about the carrier at frequenciesf_(U)=f_(c)+f_(m) and f_(L)=f_(c)−f_(m), respectively.

The upper and lower sidebands 210 and 212, respectively, will bereflected from the reference resonator with different amplitudes andphases, since they are not symmetrically placed about the center ofresonance. f_(Res). The result will be that the reflected carrier ismodulated in both phase and amplitude. Only in the special case off_(c)=f_(Res), will both the upper and lower sidebands 210 and 212 bereflected equally resulting in a reflected carrier with no amplitudemodulation at frequency f_(m). As will become apparent, the degree ofamplitude modulation at frequency f_(m), present on the reflectedcarrier signal, is an indication of the frequency error f_(c)−f_(Res).

To understand the conversion from phase to amplitude modulation,consider phasor diagrams 300 and 302 shown in FIGS. 3A-3E. FIGS. 3A-Cprovide a simplified representation of phase modulation as shown inphasor diagram 300 of a phase modulated carrier 304 such as thatimpinging on the reference resonator. Phase modulation is represented bytwo phasors 306 and 308 that counter-rotate at rates of ±2πf_(m) radiansper second at the end of the carrier phasor. When the sideband phasors306 and 308 align with the carrier 304 as shown in FIG. 3A, the phasorssum to zero. However, as shown in FIGS. 3B and 3C when the sidebandphasors 306 and 308 do not align with the carrier 304 the incident lowerand upper sidebands 310 and 312 add to the carrier 304 to advance andretard the carrier phase respectively. The result is a periodicvariation of carrier phase at the modulation frequency f_(m), with noamplitude modulation. A careful consideration of FIGS. 3A-C shows thathigher-order side bands are required to accurately describe pure phasemodulation since that the length of the carrier phasor (amplitude)changes slightly at FIGS. 3B and 3C compared to 3A; but thesimplification of ignoring higher-order side bands is sufficient toillustrate the process.

If the incident signal is a carrier 304 plus two equal amplitudesidebands 306 and 308 that add to zero as shown in FIG. 3A, thereflected sidebands 316 and 318 have different amplitudes and differentphases as shown in FIG. 3D.

Consider a reflected signal 320 in which the side bands are altered inboth amplitude 322 and phase 324 according to the frequency dependenceof the resonator reflection coefficient. As shown in FIG. 3E (not toscale) the reflected upper side band 326 is reflected with much greateramplitude than the reflected lower sideband 328. Now it is no longerpossible for the side bands to cancel in any orientation, and theirsummation follows a roughly elliptical locus 330. Thus, both the lengthand phase angle of the reflected carrier change significantly and thereis both amplitude and phase modulation, at frequency f_(m), of thereflected carrier whenever f_(c)≠f_(Res).

Furthermore, when the frequency error has the opposite sign, therelative changes in reflected sidebands 326 and 328 are swapped, and thelower side band 328 will dominate so that motion of the summation aroundthe elliptical locus 330 is reversed. The phase of the amplitudemodulation signal seen at the detector output will therefore reverserelative to the original modulating signal and the sign of the DC signalrecovered by the synchronous detector also reverses and for frequencyerrors f_(c)−f_(Res) less than the bandwidth of the resonator, thesynchronous detector output is proportional to f_(C)−f_(Res).

For an incident carrier having f_(c)≠f_(Res), the conversion of incidentphase modulation sidebands to reflected amplitude modulation sidebandsvaries with modulation frequency f_(m) in a manner proportional to

$\left. \left. {{\left. {\left\lbrack \left( \frac{dA}{df} \right. \right.❘}_{fm} \right)^{2} + \left( \frac{dP}{df} \right.}❘}_{fm} \right)^{2} \right\rbrack^{1/2}$where

${\frac{dA}{df}❘}_{fm}$is the slope of the resonator amplitude reflection coefficient, and

$\left. {\left( \frac{dP}{df} \right.❘}_{fm} \right)^{2}$is the slope of the phase reflection coefficient, at frequencyf_(Res)+f_(m). The conversion is therefore most efficient where theslopes of the curves 204 and 206 in FIG. 2B are high, that is wheref_(m) is small. The conversion factor varies relative to its maximum ina manner indicated by the dashed curve 214 in FIG. 2 .

It should be understood that the width of the dip in the amplitudereflection coefficient 204, or alternatively, the peak in the phase toamplitude conversion 214, depends on the coupling coefficient of theresonator and it is not the transmission bandwidth of the resonance. Ifthe coupling of the resonator is close to critical, the conversion fromphase to amplitude modulation will be maximal and the optimal range forf_(m) may be very much less than the resonator bandwidth. For example,in a critically coupled resonator with f_(Res)=10 GHz and a transmissionbandwidth of 400 kHz, the width of the peak in conversion from phase toamplitude, as measured by a 3 dB change from the maximum, is less than 1kHz.

Another embodiment of a frequency stabilized microwave source 400 isshown in FIG. 4 as a modification of the microwave source 100 shown inFIG. 1 with like numbers used to identify like elements. The phaseshifter 116 is eliminated and the modulation source 124 is summed via asumming node 402 with Pound-servo voltage 112 into the VCO tuning port114 to cause frequency modulation of the VCO 102. Frequency modulationand phase modulation being related simply by a factor of 1/jω, the phasemodulation of the carrier occurs with a 90° phase shift relative to themodulation source. However, this is easily accounted for in arrangingthe relative phases of the signals at synchronous detector 126.

Referring now to FIG. 5 , a frequency stabilized microwave source 500 isarranged such that a reference resonator 502 also forms part of anoscillator loop 504 including a bandpass filter (BPF) 506 tuned to passthe resonant frequency f_(Res) and a loop amplifier 510 so that aseparate VCO is not required. FIG. 5 , like FIG. 4 , achieves the phasemodulation of the signal incident on the reference resonator throughfrequency modulation.

Any phase change in oscillator loop 504 will cause a frequency change.Therefore, the frequency can be modulated by feeding a modulation signal516 from a modulation source 518 to any phase shifter in oscillator loop504. The Pound-servo feedback and the modulation signal can be appliedto a common phase shifter or separate phase shifters. If a common phaseshifter can provide the performance for both frequency modulation andtuning required by Pound-servo feedback it can be used. If the phaseshift range over which the required accuracy is achieved for modulationis insufficient to accomplish the required tuning for the Pound-servofeedback than separate phase shifters may be required.

If a common phase shifter is used, the Pound-servo voltage 520 andmodulation signal 516 are summed by summing node 522 and input to thecommon modulation phase shifter, either a loop phase shifter 508 beforethe loop amplifier 510 or a modulation phase shifter 524 after the loopamplifier 510. The other phase shifter is eliminated in thisconfiguration. It may be preferable to position the common phase shifterbefore the loop amplifier 510 in order to use the full amplifier outputpower.

If separate phase shifters are used, the Pound-servo voltage 520 isinput to loop phase shifter 508 and modulation signal 516 is input tomodulation phase shifter 524. Summing node 522 is eliminated in thisconfiguration. The positions of the phase shifters in the loop may beflipped and will be dictated by practical considerations such as canthey handle the amplified output power.

The phase modulation induced through frequency modulation in signal 512may lead to amplitude modulation in a reflected signal 530 directed by acirculator 532 to an amplitude detector 534. Any such amplitudemodulation will cause a fluctuating signal on the detector output; thefluctuation signal having the same modulation frequency f_(m). Asynchronous detector 536 converts the fluctuation in amplitude detectoroutput at modulation frequency f_(m) into a DC frequency error voltage538 representing the magnitude of the fluctuation. This DC voltage isthen amplified and preferably integrated 540 to produce the Pound-servovoltage 520 that is either fed to summing node 522 or directly to loopphase shifter 508.

As shown in FIG. 5 , a frequency-stabilized output signal 542 is takenfrom a different output 544 positioned after the reference resonator 502in the oscillator loop 504, which typically results in reduced noise inthe oscillator output due to the filtering action of the referenceresonator 502. However, the output might equally well be taken frombefore the reference resonator as in FIGS. 1 and 4 without otherwiseaffecting the function of the microwave source.

SUMMARY

The following is a summary that provides a basic understanding of someaspects of the disclosure. This summary is not intended to identify keyor critical elements of the disclosure or to delineate the scope of thedisclosure. Its sole purpose is to present some concepts of thedisclosure in a simplified form as a prelude to the more detaileddescription and the defining claims that are presented later.

The present disclosure describes the use of an IQ mixer in thePound-servo loop to detect amplitude modulation of the signal reflectedfrom the reference resonator. By properly configuring the IQ mixer sothat the LO and RF inputs are maintained in quadrature at the Q mixer,hence in-phase at the I mixer, lower levels of amplitude modulation maybe detected at lower modulation frequencies compatible with optimalchoices of resonator coupling and maximal phase to amplitude conversion.

In a frequency-stabilized microwave source that uses Pound-stabilizationor Pound-servo techniques to stabilize the oscillation frequency, theamplitude detector is implemented as an IQ mixer in which the signalreflected from the reference resonator provides the RF input and the Ioutput produces a demodulated signal at modulation frequency f_(m)representative of the amplitude modulation (AM) of the reflectedcarrier. The demodulated signal is applied to the synchronous detector.A phase error voltage indicative of a phase difference between theincident and reflected carriers provided at the Q output is integratedand fed back to a LO phase shifter where it is combined with a portionof the signal impinging on the resonator to produce a signal at the LOinput such that the LO and RF inputs are maintained in quadrature at theQ mixer, hence in-phase at the I mixer. As a result, the synchronousdetector produces a frequency error voltage proportional to thefrequency error f_(c)−f_(Res), which in turn reduces the frequency errorin the oscillator loop towards zero.

The IQ mixer so configured can be used as the AM detector in anyfrequency-stabilized microwave source that utilizes Pound-stabilizationor Pound-servo techniques. For example, those techniques may utilize areference resonator as the frequency-determining element within anoscillator loop and modulate the phase of the signal incident on theresonator by applying a modulation signal to a phase shifter at anypoint in the oscillator loop. One or more phase shifters may be providedin the oscillator loop so that the modulation signal may be combinedwith or separate from the Pound-servo feedback signal. Furthermore,those techniques may include configurations in which the referenceresonator is separate to a VCO that generates a microwave signal and thephase modulation is achieved by means of a phase shifter between the VCOoutput and reference resonator or by combining the modulation signalwith the Pound-servo feedback signal applied to tune the VCO.

In an embodiment, a quadrature compensation voltage may be summed withthe phase error voltage at the Q output at the input to the integratorto compensate for non-idealities in the quadrature phase shift betweenthe LO signals at the individual I and Q mixers.

These and other features and advantages of the disclosure will beapparent to those skilled in the art from the following detaileddescription of preferred embodiments, taken together with theaccompanying drawings, in which:

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 , as described above, is a known configuration of a frequencystabilized microwave source in which Pound-stabilization is used tostabilize an oscillator against an independent reference resonator;

FIGS. 2A-2B, as described above, are plot of the incident and reflectedspectra from the reference resonator;

FIGS. 3A-3E, as described above, are phasor diagrams illustrating theconversion from phase to amplitude modulation;

FIG. 4 , as described above, is another known configuration of afrequency stabilized microwave source in which Pound-stabilization isused to stabilize an oscillator against an independent referenceresonator;

FIG. 5 , as described above, is a known configuration of a frequencystabilized microwave source in which Pound-stabilization is implementedwith a reference resonator that also forms part of an oscillator loop;

FIG. 6 is a plot of noise floors for a diode and a mixer as AMdetectors;

FIG. 7 is a schematic diagram and symbolic representation of an IQmixer;

FIG. 8 is a schematic diagram of a core portion of a Pound-stabilizedmicrowave source that includes an IQ mixer configured to perform theamplitude modulated detection;

FIG. 9 is an alternate configuration of an IQ mixer to compensate fornon-idealities in the individual I and Q mixers;

FIG. 10 is a first embodiment of a Pound-stabilized frequency stabilizedmicrowave source in which an IQ mixer is configured to perform the AMdetection;

FIG. 11 is a second embodiment of a Pound-stabilized frequencystabilized microwave source in which an IQ mixer is configured toperform the AM detection; and

FIG. 12 is a first embodiment of a Pound-stabilized frequency stabilizedmicrowave source in which an IQ mixer is configured to perform the AMdetection.

DETAILED DESCRIPTION

The present disclosure describes the use of an IQ mixer instead of asimple diode in the Pound-servo loop to detect amplitude modulation ofthe signal reflected from the reference resonator. By properlyconfiguring the IQ mixer so that the LO and RF inputs are maintained inquadrature at the Q mixer, lower levels of amplitude modulation may bedetected at lower modulation frequencies compatible with optimal choicesof resonator coupling and maximal phase to amplitude conversion.

In a frequency-stabilized microwave source using Pound-stabilization orPound-servo techniques, a diode detector is a very simple and low-costway of recovering the amplitude modulation in the reflected signal ofthe reference resonator. However, when choosing a detector, the signalto noise ratio (SNR) needs to be considered. Diodes are non-lineardevices where the conversion ratio, that is the signal voltage producedfor a given change in input amplitude—sometimes written in mv/dB,depends upon the amplitude of the input carrier as well as the loadresistance seen by the diode as shown in FIG. 1 of Serge Grop and EnricoRubiola “Flicker Noise of Microwave Power Detectors”, 2009 IEEEInternational Frequency Control Symposium Joint with the 22nd EuropeanFrequency and Time forum, 2009, pp. 40-43. Diode detectors also producenoise which will limit the smallest detectable signal, the noise havingboth a flicker, or 1/f, component dependent on the incident carrierpower and a white floor often set by the load resistance.

Inherent within a synchronous detector is a low pass filter (LPF)element so that the synchronous detector is sensitive to only abandwidth of frequencies about the modulation frequency f_(m). Thus, anadvantage in the Pound stabilization scheme is that only the noise powerwithin the bandwidth of the synchronous detector about f_(m),contributes to noise at the diode detector output. Thus, for a diodedetector with noise 600 such as that shown by the dashed line in FIG. 6, it would be appropriate to select f_(m) greater than about 10 kHz toachieve the lowest detector noise.

When choosing a diode detector, the designer must also consider theamount of power present at the diode detector which results from thepower incident on the resonator and the resonator reflection coefficientwhich results from the resonator coupling coefficient β. If very lowpower is reflected onto the diode, as will be the case if the resonatoris critically coupled, the diode conversion ratio will be low and if ahigh load resistor is chosen to improve the conversion ratio, theresistor will contribute higher thermal noise. Alternatively, if theresonator coupling is set to a less optimal value, so that higher powerhits the diode, the conversion ratio may be higher, and a lower loadresistor may be used. However, as the power incident on the diodeincreases, the flicker, or 1/f, noise increases so that it may becomenecessary to increase the modulation frequency, f_(m), to improve signalto noise ratio. Thus, use of a diode detector dictates less than theoptimal choices dictated by maximizing the phase to amplitudeconversion, that is close to critical coupling at the resonator andsmall f_(m) as discussed previously.

Mixers may be used as phase detectors at microwave frequencies byarranging for input LO and RF signals to be in quadrature (90° relativephase), see for example Stephan R. Kurtz, “Mixers as Phase Detectors”Tech-note, Watkins-Johnson Company Vol. 5 No. 1 January/February 1978.Conversely, if the input LO and RF signals are in phase (0° relativephase), the mixer is responsive to amplitude modulation of the RFsignal. At intermediate phase angles between LO and RF a mixer will besensitive to both phase and amplitude modulation, the relativesensitivities varying as the tangent of the phase angle. Since a mixermay be arranged to detect amplitude modulation it may also be consideredas an alternative to a diode detector.

Given the wide variability of devices and operating conditions it isalways possible to find exceptions to sweeping comparisons of therelative performance of diodes and mixers as AM detectors. Generally,conversion ratios for diodes and mixers are similar, but mixer noiselevels can be significantly lower. FIG. 6 shows examples of the noisepower (noise floor) 600 and 602 for diode and mixer detectors,respectively, in a typical microwave AM detection application. The diodenoise floor has been reported, for example, in the Grop and Rubiolapaper. The mixer noise floor has been reported in CA. Barnes et. al.,“Residual PM Noise Evaluation of Radio Frequency Mixers” 2011 JointConference of the IEEE International Frequency Control and the EuropeanFrequency and Time Forum (FCS) Proceedings, 2011, pp 1-5.

The lower noise power that can be achieved using a mixer as shown by thesolid line 602 in FIG. 6 offers a significant advantage. As is evidentfrom FIG. 6 , modulation frequency f_(m) need only be greater than about100 Hz to reach the minimum mixer noise, compared to the previouslymentioned 10 kHz for the diode. This is compatible with an optimalchoice for maximal phase to amplitude conversion.

While these advantages are evident, the in-phase relationship betweenthe LO and RF signals at the I mixer must be maintained at a constantvalue close to 0° to minimize the detection of the phase modulation inthe reflected signal. If the in-phase relationship is not maintained,then contamination of the amplitude detection by unwanted detection ofattendant phase modulation may negate the advantage of lower noise powercompared to that of the diode detector. The phase difference between theLO and RF signals will drift over time and temperature, which means thedegree of contamination by phase modulation and the consequent frequencyerror drifts.

FIG. 7 is a schematic diagram 700 and symbol 702 of a circuit referredto as an IQ mixer 704. IQ mixer 704 includes a pair of conventionalmixers; an I mixer 706 and a Q mixer 708, and a circuit element 710 forproviding a fixed 90° phase shift between the LO signals at each mixer.Circuit element 710 is commonly a microwave transmission line structureor power splitter such as a branch-line coupler, also known as aquadrature hybrid, which splits the incoming common LO signal 712 at LOinput port 713 into two equal signals 714 and 716 having relative phases90° apart. An input RF signal 720 at RF input port 715 is also split bya microwave power splitter 717 into equal components 722 and 724, eachcomponent being fed to one of the mixers in the IQ arrangement. Note,equivalently the 90° shift may be imparted on the RF inputs instead ofthe LO inputs. Depending on the phase relationship between the LO and RFsignals applied to the arrangement, if the lower mixer in FIG. 7 issensitive to phase modulation (PM) then it would be identified as the“Q”, (quadrature) mixer 708, while the upper mixer is the “I” (inphase), or AM sensitive mixer 706. Q mixer 708 produces a down-convertedQ signal 730 at its Intermediate Frequency (IF) port 732 and I mixer 706produces a down-converted I output 734 at its IF port 736. IQ mixers foroperation at microwave frequencies are readily available as integratedcircuits housed in a single package. These devices therefore provide ameans of measuring both AM and PM signal modulations simultaneouslywherein only one phase adjustment between the common LO and RF signalsis required. For convenience an IQ mixer can be represented by thesimpler schematic symbol 702 shown on the right in FIG. 7 .

This disclosure relates to the method by which the amplitude modulationof the reflected signal is detected; and it may be applied in anyimplementation of frequency-stabilized microwave source usingPound-stabilization or Pound-servo techniques. This disclosure replacesthe diode detector with an IQ mixer (specially configured) in order toachieve lower AM detection noise floor at lower modulation frequencies.The IQ mixer so configured addresses and mitigates any phase drift overtime and temperature, hence any drift in phase modulation contamination.

Frequency-stabilized microwave sources that use Pound-stabilization orPound-servo techniques include an oscillator loop that provides gain>1and a phase shift of a multiple of 2pi, a modulation source configuredto modulate a microwave carrier signal at carrier frequency f_(c) atmodulation frequency f_(m) with upper and lower sidebands, a referenceresonator, having a resonant frequency f_(Res), to reflect a portion ofthe microwave signal, and a Pound-servo loop to detect differencesbetween the carrier and resonant frequencies via AM of the reflectedsignal at the modulation frequency f_(m) and feedback an error signal todrive the carrier frequency to the resonant frequency to therebystabilize the carrier signal at the resonator frequency. As describedpreviously, in different configurations the modulation can be performeddirectly in the phase domain or indirectly in the frequency domain.Furthermore, in different configurations oscillation may be provided byan arrangement wherein the resonator is separate to a VCO or anarrangement where the reference resonator is included within anoscillating loop. The present disclosure uses an IQ mixer in thePound-servo loop to detect amplitude modulation of the signal reflectedfrom the reference resonator.

Referring now to FIG. 8 , any such Pound-stabilized microwave sources inaccordance with the present disclosure include a core circuit 800 inwhich a microwave signal 802 passes through a circulator 804 and isreflected off a reference resonator 806. A modulation source 808generates a modulation signal 810 at modulation frequency f_(m) to applyphase modulation, either directly or indirectly through frequencymodulation, to microwave signal 802 impinging on the resonator. Thedegree of amplitude modulation at frequency f_(m), present on thereflected carrier signal, is an indication of the frequency errorf_(c)−f_(Res) between the carrier frequency f_(c) and the resonatorfrequency f_(Res). The reflected signal 812 is re-directed by circulator804 to an AM detector 814. A synchronous detector 816 converts thefluctuation in amplitude detector output at modulation frequency f_(m)into a DC frequency error voltage 818 representing the magnitude of thefluctuation. This DC voltage is then amplified and preferably integratedvia integrator 820 to provide Pound-servo gain to produce Pound-servovoltage 822 that is fed back (to the tuning port of the VCO or into aphase shifter within an oscillator loop) to drive the carrier frequencyf_(c) to the resonator frequency f_(Res).

In accordance with the disclosure, the AM detector 814 is implemented asan IQ mixer 830 to detect amplitude modulation (AM) in the reflectedsignal 812. This achieves the lower noise floor 602 associated with amixer shown in FIG. 6 and thus facilitates the use of lower modulationfrequencies f_(m) (e.g., f_(m) need only be greater than about 100 Hz toreach the minimum mixer noise), which is compatible with optimal choicesof resonator coupling and maximal phase to amplitude conversion.Furthermore, feedback is provided from the output of the Q mixer to theLO input of the IQ mixer to hold the phase at the LO input to the Qmixer at approximately 90 degrees such that the phase at the LO input tothe I mixer is approximately 0 degrees. This is also referred to as“maintaining the LO and RF inputs in quadrature at the Q mixer.” This iscritical to minimize the detection of any phase modulation in thereflected signal 812 such that the detected signal at the output of theI mixer is solely attributed to amplitude modulation in the reflectedsignal. If the LO input to the Q mixer is not maintained at 90 degrees,phase modulation in the reflected signal 812 will induce an error in thePound-servo voltage 822.

IQ mixer 830 includes an I mixer 832 having LO and RF input ports 834and 836, respectively, and an IF output port 840 and a Q mixer 842having LO and RF input ports 844 and 846, respectively, and an IF outputport 850. A microwave power splitter 852 splits a LO signal 854 intofirst and second LO signals 856 and 858, respectively, having relativephases 90° apart, that are applied to LO input ports 834 and 844,respectively. A microwave power splitter 860 splits the reflected signal812 into first and second RF signals 862 and 864, respectively, havingthe same phase, that are applied to RF input ports 836 and 846,respectively. The I and Q mixers produce I and Q down-converted signals870 and 872, respectively, at IF outputs ports 840 and 850,respectively.

In operation, a portion 880 of the signal 802 directed towards thereference resonator 806 is coupled off and sent via a local oscillator(LO) phase shifter 882 as LO signal 854 to the LO port of the IQ mixer.For the purposes of explanation, assume that the voltage applied to theLO Phase shifter 882 is at zero volts and that suitable transmissionline path lengths, or other phase-shifting means such as combinations oflumped circuit elements, not shown in the figures, have been provided sothat the LO and RF signals 854 and 812, respectively, arrive at theinput ports to the IQ mixer with some small phase difference, Δφ≠0.Practical phase shifters will typically function with only one polarityof control voltage, but it is a simple matter to sum in an offset tomeet this condition.

The I mixer 832 in this scenario, while being predominantly responsiveto amplitude modulation of the reflected signal 812, also has a smallsensitivity to phase modulation in the reflected signal so that the DCfrequency error voltage 818 at the synchronous detector output can beconsidered as a summation of two component voltages, V_(AM) and V_(PH)resulting from amplitude and phase modulations of the reflected carrier.The phase modulation applied to the carrier and reflected from theresonator (still as phase modulation) will therefore result in a smallV_(PH)=V_(Error). In response to V_(Error), the Pound-servo voltage 822will adjust the frequency of the oscillator loop where theamplitude-to-phase conversion action of the resonator reflectionproduces sufficient amplitude modulation to provide V_(AM)=−V_(Error) sothat the synchronous detector output goes to zero volts and theoscillator loop remains at a frequency f_(c)=f_(Res)+f_(Error).

Now, the small phase error, Δφ, may be considered as a “DC modulation”of the phase between LO and RF signals 858 and 864 at the input to the Qmixer 842. The down-converted Q signal 872, which is a phase errorvoltage, will therefore carry a DC signal, Vφ, proportional to Δφ. It isimportant to note that Vφ results from the phase difference between thecarriers (incident and reflected) at the mixer, not from the modulationside bands. The sidebands will produce a fluctuating signal at the Qoutput having frequency f_(m).

If Vφ is amplified and fed back via a LPF or amplifying integrator 890to the LO phase shifter 882 with the correct sign, then the phase errorΔφ will be reduced, simultaneously adjusting the Q mixer 842 towardspure phase sensitivity and the I mixer 832 towards pure amplitudesensitivity. As the I mixer 832 loses phase sensitivity the voltageV_(Error) diminishes with a consequence that the frequency errorf_(Error) in the Pound-servo is reduced.

The fluctuating signal at the Q output resulting from the phasemodulation of the carrier is effectively removed from consideration bythe low pass filtering action of the integrator (or LPF). It will beapparent that in the preceding discussion a simple low pass filter andamplifier between the Q output 850 and LO Phase shifter 882 would serveto produce a similar result. However, the infinite DC gain provided byan integrator will result in the phase error Δφ diminishing to zero. Thetime constant of the integrator in the LO Phase feedback loop can belong since it only needs to compensate for relatively slow thermal oraging effects, and it is therefore highly effective at removing themodulation signal at f_(m).

As a result, the down-converted I signal 870 includes a demodulationsignal having frequency f_(m). The demodulation signal has a magnitudeand sign representative of the amplitude modulation of the reflectedcarrier. As such the synchronous detector produces frequency errorvoltage 818 proportional only to the frequency error f_(c)−f_(Res),which in turn reduces the frequency error in the oscillator loop towardszero.

The discussion so far has assumed that the independent I and Q mixerswithin the IQ mixer are ideally responsive as if the phase differencebetween their LO signals was exactly 90° (quadrature). This ideal statemay not be realized in practice and there will usually be some smallquadrature error. As a result, even if the LO Phase shifter is adjustedso that Q mixer is perfectly phase sensitive, the I mixer will retainsome slight sensitivity to phase modulation, resulting in aphase-modulation induced frequency error, f_(PMError).≠0, in thePound-servo. Diode detectors are also slightly sensitive to phasemodulation. Fundamentally, both a diode detector and an AM-sensitivemixer will have some low-pass bandwidth which is responsible forrejecting the carrier frequency f_(c) at the detector output. Theminimum detector bandwidth and therefore maximum rejection of f_(c) isgoverned by the need to pass the modulation frequency f_(m). Therefore,both diode and mixer amplitude detectors become subject to the samefundamental limit on rejection of the phase modulation signal and thesame minimum f_(PMError).

In the case of the arrangement described in FIG. 8 , the realization ofthis fundamental limit may require compensation for any small quadratureerror in the IQ mixer. This compensation is achieved by the addition ofa small quadrature compensation voltage 900 via summing node 902 at theinput to the integrator 890 as shown in FIG. 9 . Since the sign off_(Error) resulting from a quadrature error in the IQ mixer 830 reversesas the quadrature error passes through zero, it is a simple matter todetermine the magnitude of the required quadrature compensation voltage900 by adjusting it to reduce sensitivity of the demodulation signal tophase modulation in the reflected signal.

In any case, a requirement that the carrier frequency fc remainsconstant does not necessarily extend to requiring that f_(PMError)=0,only that, f_(PMError)=constant. The stability of f_(PMError) is aseparate consideration and does not detract from the ability of themixer amplitude detector to detect a smaller frequency deviation in thePound-servo loop because of its lower noise floor.

Thus, it is demonstrated that the disclosed technique maintains thecorrect phase relationship between the LO and RF signals at the IQ mixersuch that the Q output is maintained maximally sensitive to amplitudemodulation and minimally sensitive to phase modulation, thereby enablingpractical use of a mixer instead of a diode as the detector in aPound-stabilization system and realization of the advantages of themixer's lower noise floor.

As described, the IQ mixer can be implemented in any Pound-stabilized orPound-servo frequency-stabilized microwave source. FIGS. 10-12illustrate three of the known Pound-stabilized microwave sourcetopologies illustrated in FIGS. 1, 4 and 5 , respectively.

Referring now to FIG. 10 , a frequency stabilized microwave source 1000uses the Pound-stabilization technique to stabilize a microwaveoscillator i.e., a VCO 1002 against a reference resonator 1004 separateto the VCO to generate a frequency-stabilized output signal 1006 atoutput port 1008. The VCO generates an output signal 1010, a portion ofwhich is taken as frequency-stabilized output signal 1006. The signalfrequency can be tuned by a Pound-servo voltage 1012 applied to theVCO's tuning port 1014. Output signal 1010 is passed through a phaseshifter 1016 and a circulator 1018 before impinging on referenceresonator 1004. The frequency of the VCO 1002 must be tunable to matchthe center frequency of the resonator, f_(Res). Typically, the resonatormay be a cavity resonator which may include a dielectric element,although the technique may be applied using any resonator with asuitable center frequency and stability.

A portion of the output signal 1010 impinging on the resonator will bereflected in accordance with well-known properties of resonators asreflected signal 1020. This reflected energy is directed by circulator1018 onto an amplitude detector 1022 which is implemented as an I/Qmixer 1023. A modulation source 1024 generates a modulation signal 1025at a modulation frequency, f_(m) to apply phase modulation to outputsignal 1010 impinging on the resonator by way of the phase shifter 1016.The phase modulation may lead to amplitude modulation in the reflectedsignal 1020 seen by the amplitude detector 1022. Any such amplitudemodulation will cause a fluctuating signal on the detector output, saidfluctuation having the same modulation frequency f_(m). IQ mixer 1023 isconfigured to minimize or eliminate any contribution to the demodulationsignal 1032 caused by phase modulation in the reflected signal 1020.

IQ mixer 1023 receives reflected signal 1020 at its RF input 1030 andproduces a demodulation signal 1032 at modulation frequency f_(m)representative of the AM of the reflected carrier at its I output 1034.An LO phase shifter 1042 receives a portion 1044 of the signal 1010. Aphase error voltage 1036 indicative of a phase difference between theincident and reflected carriers is provided at the mixer's Q output1038, which is integrated 1040 and fed back to the LO phase shifter 1042as an approximately DC signal where it modifies the effective phaseshift through LO phase shifter 1042 to produce a signal 1046 at themixer's LO input 1048 such that the LO and RF inputs are maintained inquadrature at the Q mixer, hence in-phase at the I mixer.

A synchronous detector 1050 converts demodulation signal 1032 into a DCfrequency error voltage 1052 representing the magnitude of thefluctuation. By maintaining the inputs to the Q mixer in quadrature, thesynchronous detector produces a frequency error voltage proportionalonly to any amplitude modulation of the reflected signal and thereby thefrequency error f_(c)−f_(Res), which in turn reduces the frequency errorin the oscillator loop towards zero. This DC voltage is then amplifiedand preferably integrated via integrator 1054 to provide Pound-servogain to produce Pound-servo voltage 1012 that is fed to the tuning port1014 of VCO 1002.

Another embodiment of a frequency stabilized microwave source 1100 isshown in FIG. 11 as a modification of the microwave source 1000 shown inFIG. 10 with like numbers used to identify like elements. The phaseshifter 1016 is eliminated and the modulation source 1024 is summed viaa summing node 1102 with Pound-servo voltage 1012 into the VCO tuningport 1014 to cause frequency modulation of the VCO 1002. Frequencymodulation and phase modulation being related simply by a factor of1/jω, the phase modulation of the carrier occurs with a 900 phase shiftrelative to the modulation source. However, this is easily accounted forin arranging the relative phases of the signals at synchronous detector1050. The IQ mixer 1023 is configured and operates as previouslydescribed.

Referring now to FIG. 12 , a frequency stabilized microwave source 1200is arranged such that a reference resonator 1202 also forms part of anoscillator loop 1204 including a bandpass filter (BPF) 1206 tuned topass the resonant frequency f_(Res) and a loop amplifier 1210 so that aseparate VCO is not required. FIG. 12 , like FIG. 11 , achieves thephase modulation of the signal incident on the reference resonatorthrough frequency modulation.

Any phase change in oscillator loop 1204 will cause a frequency change.Therefore, the frequency can be modulated by feeding a modulation signal1216 from a modulation source 1218 to any phase shifter in oscillatorloop 1204. The Pound-servo feedback and the modulation signal can beapplied to a common phase shifter or separate phase shifters positionedanywhere in oscillator loop 1204. If a common phase shifter can providethe performance for both frequency modulation and tuning required byPound-servo feedback it can be used. If the phase shift range over whichthe required accuracy is achieved for modulation is insufficient toaccomplish the required tuning for the Pound-servo feedback thanseparate phase shifters may be required. As shown, a common phaseshifter 1224 is positioned at the output of loop amplifier 1210. APound-servo voltage 1220 and modulation signal 1216 are summed bysumming node 1222 and input to the common phase shifter 1224.

The phase modulation induced through frequency modulation of signal 1212may lead to amplitude modulation in a reflected signal 1230 directed bya circulator 1232 to an amplitude detector 1234 implemented as an IQmixer 1236 in which the phase error voltage at the Q output isintegrated 1238 and summed with a portion of signal 1212 by a LO phaseshifter 1240 and feedback to the LO input to hold the LO and RF inputsof the Q mixer in quadrature. Any such amplitude modulation will cause afluctuating signal on the detector output; the fluctuating signal havingthe same modulation frequency f_(m). IQ mixer 1236 is configured tominimize or eliminate any contribution to a demodulation signal 1241 atthe I output caused by phase modulation in the reflected signal 1230.

A synchronous detector 1242 converts the fluctuation in the I outputsignal at modulation frequency f_(m) into a DC frequency error voltage1244 representing the magnitude of the fluctuation. By maintaining theinputs to the Q mixer in quadrature, the synchronous detector produces afrequency error voltage proportional only to any amplitude modulation ofthe reflected signal and thereby the frequency error f_(c)−f_(Res),which in turn reduces the frequency error in the oscillator loop towardszero. This DC voltage is then amplified and preferably integrated viaintegrator 1246 to produce the Pound-servo voltage 1220 that is fed tosumming node 1222.

A frequency-stabilized output signal 1250 is taken from a differentoutput 1242 positioned after the reference resonator 1202 in theoscillator loop 1204, which typically results in reduced noise in theoscillator output due to the filtering action of the reference resonator1202. However, the output might equally well be taken from before thereference resonator without affecting the function of the microwavesource.

While several illustrative embodiments of the disclosure have been shownand described, numerous variations and alternate embodiments will occurto those skilled in the art. Such variations and alternate embodimentsare contemplated, and can be made without departing from the spirit andscope of the disclosure as defined in the appended claims.

We claim:
 1. A frequency-stabilized microwave source, comprising: anoscillator configured to produce a signal including a carrier at amicrowave carrier frequency fe and having a tuning port responsive to atuning voltage to control the frequency fe; a Pound-stabilization loopcomprising: a modulation source configured to generate a modulationsignal at modulation frequency fm, said modulation signal configured toadd upper and lower sidebands at fc+fm and fc−fm to the signal; areference resonator having a center frequency fRes configured to receiveat least a portion of the signal and to produce a reflected carrier andreflected upper and lower sidebands, a portion of the reflected upperand lower side bands being representative of amplitude modulation (AM)of the reflected carrier at modulation frequency fm and an indicator ofa frequency error fe−fRes; an AM detector responsive to a reflectedsignal to produce a demodulation signal representative of the amplitudemodulation of the reflected carrier at modulation frequency fm; and asynchronous detector that mixes the demodulation signal with themodulation signal to produce a frequency error voltage indicative of thefrequency error fe−fRes, said frequency error voltage controlling thetuning voltage to reduce the frequency error towards zero, wherein theAM detector comprises an IQ mixer having LO and RF input ports and I andQ output ports, wherein the RF input port receives the reflected signaland the demodulation signal is produced at the I output port, whereinthe IQ mixer produces a phase error voltage indicative of a phasedifference between the carrier and the reflected carrier at the Q outputport, an LO phase shifter responsive to the phase error voltage, thatsamples the signal and produces a phase-shifted signal, wherein thephase-shifted signal is applied to the LO input port of the IQ mixersuch that the demodulation signal produced at the I output port isinsensitive to phase modulation in the reflected signal.
 2. Thefrequency-stabilized microwave source of claim 1, wherein the oscillatorcomprises a voltage-controlled oscillator (VCO) that generates thesignal, wherein the reference resonator is separate from the VCO.
 3. Thefrequency-stabilized microwave source of claim 2, further comprising: aphase shifter at the output of the VCO, said phase shifter responsive tothe modulation signal to apply phase modulation to the signal to add theupper and lower sidebands.
 4. The frequency-stabilized microwave sourceof claim 2, further comprising: a summing node at the tuning port of theVCO, said summing node responsive to the modulation signal and thefrequency error voltage to frequency modulate carrier frequency f_(c).5. The frequency-stabilized microwave source of claim 2, furthercomprising: an integrator configured to integrate the frequency errorvoltage that is fed to the tuning port of the VCO.
 6. Thefrequency-stabilized microwave source of claim 1, wherein the oscillatorincludes the reference resonator as a frequency-determining elementwithin an oscillator loop, further comprising a first phase shifter inthe oscillator loop, said phase shifter having a voltage-control portbeing responsive to the tuning voltage to control the frequency f_(c).7. The frequency-stabilized microwave source of claim 6, furthercomprising: a second phase shifter in the oscillator loop responsive tothe modulation signal to apply phase modulation to the signal to add theupper and lower sidebands.
 8. The frequency-stabilized microwave sourceof claim 7, wherein the first and second phase shifters are a commonphase shifter, further comprising a summing node that sums themodulation signal and the frequency error voltage.
 9. Thefrequency-stabilized microwave source of claim 8, further comprising: anintegrator configured to integrate the frequency error voltage that isprovided to the summing node.
 10. The frequency-stabilized microwavesource of claim 1, further comprising a lowpass filter (LPF) orintegrator between the Q output port and the LO phase shifter.
 11. Thefrequency-stabilized microwave source of claim 10, further comprising: asumming node at the input to the LPF or integrator, said summing nodeconfigured to sum the phase error voltage at the Q output port and aquadrature compensation voltage having a magnitude set to reducesensitivity of the demodulation signal to phase modulation in thereflected signal.
 12. The frequency-stabilized microwave source of claim1, wherein the IQ mixer comprises: an I mixer having an LO input port,an RF input port and the I output port; a Q mixer having an LO inputport, an RF input port and the Q output port; a first microwave powersplitter between the IQ mixer's LO input port and the LO input ports ofthe I and Q mixers that receives the phase-shifted signal and producesfirst and second LO signals; a second microwave power splitter betweenthe IQ mixer's RF input port and the RF input ports of the I and Qmixers that receives the reflected signal and produces first and secondRF signals having the same phase; wherein the first and second microwavepower splitters are arranged to produce first and second relative phasesbetween the first LO signal and first RF signal and second LO signal andsecond RF signal, respectively, the difference between first and secondrelative phases being 900, and wherein the phase-shifted signal isapplied to the LO input port of the IQ mixer such that the second LOsignal and the second RF signal at the LO input port and RF input portof the Q mixer, respectively, are maintained in quadrature such that thedemodulation signal produced at the I output port is insensitive tophase modulation in the reflected signal.
 13. The frequency-stabilizedmicrowave source of claim 1, wherein the 5 Hz<f_(m)<5 KHz.
 14. Afrequency-stabilized microwave source, comprising: an oscillatorconfigured to produce a signal including a carrier at a microwavecarrier frequency fe and having a tuning port responsive to a tuningvoltage to control the frequency fc; a Pound-stabilization loopcomprising: a modulation source configured to generate a modulationsignal at modulation frequency fm, said modulation signal configured toadd upper and lower sidebands at fc+fm and fc−fm to the signal; areference resonator having a center frequency fRes configured to receiveat least a portion of the signal and to produce a reflected carrier andreflected upper and lower sidebands, a portion of the reflected upperand lower sidebands being representative of amplitude modulation (AM) ofthe reflected carrier at modulation frequency fm and an indicator of afrequency error fc−fRes; an AM detector responsive to a reflected signalto produce a demodulation signal representative of the amplitudemodulation of the reflected carrier at modulation frequency fm; and asynchronous detector that mixes the demodulation signal with themodulation signal to produce a frequency error voltage indicative of thefrequency error fc−fRes, said frequency error voltage controlling thetuning voltage to reduce the frequency error towards zero, wherein theAM detector comprises an IQ mixer comprising: an I mixer having an LOinput port, an RF input port and an I output port; a Q mixer having anLO input port, an RF input port and a Q output port; a first microwavepower splitter that receives a phase-shifted signal and produces firstand second LO signals at the I and Q mixer input ports, respectively; asecond microwave power splitter that receives the reflected signal andproduces first and second RF signals at the I and Q mixer RF ports,respectively; wherein the first and second microwave power splitters arearranged to produce first and second relative phases between the firstLO signal and first RF signal and second LO signal and second RF signalrespectively, the difference between first and second relative phasesbeing 900; wherein the I mixer produces the demodulation signal at its Ioutput port; wherein the Q mixer produces a phase error voltageindicative of a phase difference between the carrier and the reflectedcarrier at the Q output port, an LO phase shifter responsive to thephase error voltage that samples the signal and produces thephase-shifted signal, wherein the phase-shifted signal is applied to theLO input port of the Q mixer such that the second LO signal and thesecond RF signal at the LO input port and RF input port of the Q mixer,respectively are maintained I quadrature such that the demodulationsignal produced at the I output port is insensitive to phase modulationin the reflected signal.
 15. The frequency-stabilized microwave sourceof claim 14, wherein the oscillator comprises a voltage-controlledoscillator (VCO) that generates the signal, wherein the referenceresonator is separate from the VCO.
 16. The frequency-stabilizedmicrowave source of claim 14, wherein the oscillator includes thereference resonator as a frequency-determining element within anoscillator loop, further comprising a first phase shifter in theoscillator loop, said phase shifter having a voltage-control port beingresponsive to the tuning voltage to control the carrier frequency f_(c).17. The frequency-stabilized microwave source of claim 14, furthercomprising a lowpass filter (LPF) or integrator between the Q outputport and the LO phase shifter.
 18. The frequency-stabilized microwavesource of claim 14, further comprising: a summing node at the input tothe LPF or integrator, said summing node configured to sum the phaseerror voltage at the Q output port and a quadrature compensation voltagehaving a magnitude set to reduce sensitivity of the demodulation signalto phase modulation in the reflected signal.
 19. An amplitude modulation(AM) detector for a Pound-stabilized microwave source in which areference resonator having a center frequency f_(Res) is configured toreflect a signal including a carrier at carrier frequency f_(c) andupper and lower sidebands at f_(c)+f_(m) and f_(c)−f_(m) at modulationfrequency f_(m), the AM detector comprising: an IQ mixer having LO andRF input ports and I and Q output ports; wherein the RF input port isconfigured to receive a reflected signal including a reflected carrierand reflected upper and lower sidebands, a portion of the reflectedupper and lower sidebands being representative of AM of the reflectedcarrier at modulation frequency f_(m) and an indicator of a frequencyerror f_(c)−f_(Res); wherein the IQ mixer is configured to produce ademodulation signal representative of the amplitude modulation of thereflected carrier at modulation frequency f_(m) at the I output port;wherein the IQ mixer is configured to produce a phase error voltageindicative of a phase difference between the carrier and the reflectedcarrier at the Q output port, an LO phase shifter responsive to thephase error voltage, that samples the signal and produces aphase-shifted signal, wherein the phase-shifted signal is applied to theLO input port of the IQ mixer such that the demodulation signal producedat the I output port is insensitive to phase modulation in the reflectedsignal.
 20. The frequency-stabilized microwave source of claim 19,wherein the IQ mixer comprises: an I mixer having an LO input port, anRF input port and the I output port; a Q mixer having an LO input port,an RF input port and the Q output port; a first microwave power splitterbetween the IQ mixer's LO input port and the LO input ports of the I andQ mixers that receives the phase-shifted signal and produces first andsecond LO signals; a second microwave power splitter between the IQmixer's RF input port and the RF input ports of the I and Q mixers thatreceives the reflected signal and produces first and second RF signals;wherein the first and second microwave power splitters are arranged toproduce first and second relative phases between the first LO signal andfirst RF signal and second LO signal and second RF signal respectively,the difference between first and second relative phases being 900,wherein the phase-shifted signal is applied to the LO input port of theIQ mixer such that the second LO signal and the second RF signal at theLO input port and RF input port of the Q mixer, respectively, aremaintained in quadrature such that the demodulation signal produced atthe I output port is insensitive to phase modulation in the reflectedsignal.
 21. The frequency-stabilized microwave source of claim 19,further comprising a lowpass filter (LPF) or integrator between the Qoutput port and the LO phase shifter.